Search results

The paper aims to present the comparison between the performances of the exterior‐rotor permanent magnet synchronous motors with distributed windings and the performances of the exterior‐rotor permanent magnet synchronous motors with concentrated windings.

Finite element method is used for motors performance determination. The BLDC operation mode for the motors with different slot and pole number combination and concentrated windings was accounted for in the comparison.

In the BLDC operation mode motor structures with concentrated windings with similar slot and pole numbers exhibit at the same current density similar or even higher torque capability and lower electromagnetic torque ripple in comparison to the motor structure with distributed windings. Motor structures with 9‐slot/8‐pole, 9‐slot/10‐pole, 12‐slot/10‐pole slot and pole number combinations are the most appropriate for the BLDC operation.

The paper shows which motor structures with distributed or concentrated windings in the BLDC operation mode produce lower torque ripple and higher average torque per ampere.

The investigation was aimed at magnetically‐nonlinear dynamic model of a single‐phase transformer, where the effects of dynamic hysteresis losses are accounted for by a simplified model. Such a modelling could be applied when analyzing the transient operating conditions or the impact of nonlinear and unbalanced loads on the transformer operation and the big power systems modelling.

Secondly, an inverse form of the Jiles‐Atherton hysteresis model was applied for the hysteresis losses of a transformer defining. In that sense this paper compares and evaluates both hysteresis models, where the possible errors caused by simplified model application are exposed.

The Jiles‐Atherton model can be applied when more accurate hysteresis models are required, however, at the cost of increased model complexity and required computational effort. Apart from that the main drawback is impossible application of such a modelling, when some of the input parameters are unknown. On the other hand the simplified hysteresis model does not increase the required computational effort substantially.

Both methods have been modified in such a way that they can be used when the magnetizing curve of the iron‐core material is not available, whilst the magnetically‐nonlinear characteristic of the entire device can be determined experimentally. The aforementioned characteristic can be given in the form of an approximation polynomial or in the form of a look‐up table.

The majority of three‐phase dynamic transformer models used in commercially available electric power system transient simulation programs offer only saturated three‐phase transformer models built from three single‐phase transformer models. This paper sets out to deal with the modelling and transient analysis of a saturated three‐limb core‐type transformer.

Three iron core models I‐III are given by the current‐dependent characteristics of flux linkages. In the first model, these characteristics are given by a set of piecewise linear functions, which include saturation. In the second model, the piecewise linear functions are replaced by the measured nonlinear characteristic. The more complex third model is given by a set of measured flux linkage characteristics.

The behaviour of transformers used in electric power applications depends considerably on the properties of magnetically nonlinear iron core. The best agreement between the calculated and measured results is obtained by use of the most complex iron core model III, which takes into account magnetic cross‐couplings between different limbs, caused by saturation.

Measurement of the current‐dependent flux linkage characteristics of the 0.4 kV, 3.5 kVA laboratory transformer requires corresponding excitation of windings by three independent linear amplifiers. Current‐dependent flux linkage characteristics of the larger power transformer can be determined either by similar measurement with linear amplifiers of an appropriate power or by extracting them from the calculated magnetic field, which is done by the finite element method.

A three‐phase dynamic transformer model with the obtained iron core model III is suitable for the numerical analysis of nonsymmetric transient states in power systems.

This paper presents a three‐phase dynamic transformer model with an original iron core model III, which accounts for magnetic cross‐couplings between different limbs, caused by saturation.

The finite element (FE) method calculations are used to improve dynamic behavior of the two‐axis linear synchronous reluctance motor (LSRM) model, which is appropriate for the control design, the real time applications and the low speed servo applications. By the FE method, calculated current and position dependent flux linkages, their partial derivatives and motor thrust are approximated by the continuous functions and incorporated into the dynamic LSRM model as a nonlinear iron core model. The agreement between the calculated and the measured flux linkages, their partial derivatives and the motor thrust is very good. The agreement between all trajectories calculated by the improved dynamic LSRM model and measured during the experiment in the case of kinematic control is very good as well.

Many authors reported the decrease of performances when electric machines and electromagnetic devices were supplied by pulse width modulated (PWM) voltages. However, these statements are rarely supported by measurements performed under fair conditions. The aim of this paper is to compare the performances of a single‐phase transformer and a three‐phase permanent magnet synchronous motor (PMSM) supplied by sinusoidal and PWM voltages and to find a way to evaluate the decrease of performances when PWM voltages are applied.

In order to perform a fair comparison between performances of the tested objects supplied by sinusoidal and PWM voltages, an experimental system was built. It contains a single‐phase and a three‐phase linear rectifier for supply with sinusoidal voltages and an H‐bridge inverter and a three‐phase inverter for supply with PWM voltages. The tests and measurements were performed on a single‐phase transformer and three‐phase PMSM, where different constant loads and different modulation frequencies were used. The test conditions were identical for the supply by sinusoidal and PWM voltages. The measured data, used for the evaluation of performances, were the input and output power and the time behaviours of currents and voltages together with their THDs.

The results presented in the paper clearly show that the efficiency of the singe‐phase transformer and three‐phase PMSM decreases with the increasing level of voltage THD. To properly determine the THD of PWM voltage, the sampling frequencies above 1 MHz and special equipment are normally required. However, if the modulation frequency is not too high, also the current THD, which can be easily determined, can be used to evaluate the decrease of efficiency in the case of supply by PWM voltages.

The results presented in the paper clearly show that the efficiency of the singe‐phase transformer and three‐phase PMSM decreases with the increasing level of voltage THD.

The dynamic model of radial active magnetic bearings, which is based on the current and position dependent partial derivatives of flux linkages and radial force characteristics, is determined using the finite element method. In this way, magnetic nonlinearities and cross‐coupling effects are considered more completely than in similar dynamic models. The presented results show that magnetic nonlinearities and cross‐coupling effects can change the electromotive forces considerably. These disturbing effects have been determined and can be incorporated into the real‐time realization of nonlinear control in order to achieve cross‐coupling compensations.

The purpose of this paper is to analyze the impact of magnetic saturation on the steady‐state operation of the induction motor (IM) drive in regard to rotor field‐oriented control (RFOC). The aim of the presented two methods is to obtain the required steady‐state torque with minimal stator current, which thus reduces stator coper losses considerably.

The first method is based on an analytic calculation of the peak torque‐per‐ampere ratio curve of saturated IM. The torque characteristics obtained at a constant stator current are used to calculate that value of magnetizing current which gives the minimal stator current for the required load torque. The second method directly searches the minimal stator current for the required load torque. Experiments completely confirm the efficiency of the proposed selection of a magnetizing current reference.

Operation of the IM drive strongly depends on a proper selection of the rotor flux linkage reference value, the selection of which represents an additional degree of freedom in control design. Therefore, it can be used to optimize some of those drive features subjected to voltage and current constraints. The proposed calculation procedure is simple so that can be easily implemented in practically application. However, some additional IM data like magnetizing curve, inertia moment, and coefficient of viscous friction are necessary.

The substantial impact of saturation on the stead‐state torque characteristics of IM, determined for the constant stator current and the constant d‐axis stator current, is determined analytically and numerically.

So far the proposed analytical methods for calculation of copper losses are rather simplified and do not include the time component in the basic partial differential equations, which describe current density distribution. Moreover, when the physical parameters of the transformer (wire dimensions) are out of the certain range, the current density distribution approaches infinity. The purpose of this paper is to offer a generally applicable analytical method. The main goal of the proposed modification of the solution to the current density is improvement of the accuracy and stability of the analytical results.

This paper deals with the calculation of copper losses with various methods, which are based on a time‐dependent electromagnetic field. Analytical method is based on Maxwell equations and Helmholtz equation. Numerical calculation is performed with finite element method (FEM).

Analytical method is a very accurate and it gives results, which are very similar to the actual behaviour of the current density in the winding. However, the FEM analysis is easier to comprehend, but yet very dependent on input parameters.

The numerical analysis may not be accurate enough, because of the problems with the oscillation of the output welding current amplitude. To calculate copper losses correctly, the output welding current must be equal in all test cases, especially during the measurements.

When the physical properties exceed a certain range, the copper losses of the analyzed welding transformer cannot be calculated with existing analytical methods. The new analytical approach gives a far more realistic solution to the current density distribution and improves the accuracy and stability of the results.